The present invention relates to a tracking error signal generation circuit for an optical disk playback apparatus for playing back an optical disk and, more particularly, to a tracking error balance adjustment circuit. The present invention is applied to an audio compact disk (CD) player, the CD-ROM playback apparatus of a computer system, and the like.
The present invention also relates to a current control circuit, variable gain amplifier circuit using the same, and optical disk playback apparatus and, more particularly, to a current control circuit for supplying a control current to a gm amplifier whose transconductance gm is changed by controlling the collector current of a differential amplifier transistor. The present invention is applied to an audio compact disk (CD) player, the CD-ROM drive of a computer system, and the like.
Generally, in tracking control in a CD-ROM playback apparatus, the positional shift between a track on a rotating disk and the irradiation position of a laser beam emitted by an optical pickup is detected to control the position of the optical pickup so as to accurately irradiate the track with the laser beam.
FIGS. 1A to 1C show different examples of the irradiation positions of laser beams (main beam M and two sub-beams E and F) on a track T of a disk.
The irradiation positions of the two sub-beams E and F are set to slightly shift from each other in opposite directions along the track width on the two sides of the irradiation position of the main beam M along the track.
As shown in FIG. 1D, the main beam M is photoelectrically converted by four photodiodes A, B, C, and D in correspondence with four divided regions of the main beam M. Outputs from the four photodiodes are input to a head amplifier where the outputs are added.
Each of the two sub-beams E and F is photoelectrically converted by one photodiode. An output (E or F signal) from the photodiode is input to the tracking error signal generation circuit of the head amplifier.
This tracking error signal generation circuit removes the high-frequency components of the E and F signals, differentially adds the resultant E and F signals, and outputs the sum as a tracking error signal. Tracking servo control reduces the tracking error signal to 0.
FIG. 2 shows a conventional tracking error signal generation circuit.
A first current-to-voltage conversion circuit 71 receives an output (RF signal) from a first photodiode arranged in correspondence with one (first sub-beam E) of the two sub-beams, converts a current to a voltage, and removes a high-frequency component.
A second current-to-voltage conversion circuit 72 receives an output (RF signal) from a second photodiode arranged in correspondence with the other one (second sub-beam F) of the two sub-beams, converts a current to a voltage, and removes a high-frequency component.
Output voltages from the two current-to-voltage conversion circuits 71 and 72 are adjusted by a tracking error balance adjustment circuit 80, and differentially input to an adder circuit 73 via resistive elements R2. An output from the adder circuit 73 is input to a gain adjustment circuit 74 via a resistive element R3.
In the adder circuit 73, a feedback resistive element Rf and a capacitive element Cf for passing a high-frequency component are parallel-connected between the (xe2x88x92) input terminal and output terminal of an operational amplifier circuit OA3. A resistive element Ro and a capacitive element Co for removing a high-frequency component are parallel-connected between a reference potential and the (+) input terminal of the operational amplifier circuit OA3.
In the gain adjustment circuit 74, a feedback resistor Ru adjustable by the user is connected (e.g., externally connected to an LSI) between the (xe2x88x92) input terminal and output terminal of an operational amplifier circuit OA4. A resistive element R4 is connected between the reference potential and the (+) input terminal of the operational amplifier circuit OA4.
If a tracking error signal TE output from the gain adjustment circuit 74 is 0, the two sub-beams E and F are at ideal irradiation positions; and if the tracking error signal TE shifts in a (+) or (xe2x88x92) direction, the two sub-beams E and F shift from ideal irradiation positions to one side along the track width.
The role of the tracking error balance adjustment circuit 80 will be explained.
Even if the irradiation position of a laser beam on the track of a disk is accurate as in a case wherein tracking control is accurately done, characteristics may vary between the first system including the first sub-beam E and corresponding first photodiode and the second system including the second sub-beam F and corresponding second photodiode. In this case, an output (E signal) from the first photodiode and an output (F signal) from the second photodiode exhibit different amplitudes. To correct such variations, tracking error balance adjustment must be performed. Thus, output voltages OE and OF from the two current-to-voltage conversion circuits 71 and 72 are adjusted by the tracking error balance adjustment circuit 80, and then differentially added by the adder circuit 73. As a control signal input to the adjustment circuit 80, a DC tracking error balance adjustment voltage TEB generated by another means is input.
The tracking error balance adjustment circuit 80 comprises two current-controlled variable gain control circuits in correspondence with the voltage OE prepared by converting the E signal by the first current-to-voltage conversion circuit 71, and the voltage OF prepared by converting the F signal by the second current-to-voltage conversion circuit 72.
The gains of the two variable gain control circuits are controlled by an output current (control current) prepared by converting an input tracking error balance adjustment voltage TEB into a current by a current control circuit 90. In general, the two circuits are differentially controlled to increase the gain of one circuit and decrease the gain of the other circuit.
The conventional tracking error balance adjustment circuit 80 shown in FIG. 2 will be explained.
The input voltages OE and OF of these circuits are respectively input to current-controlled variable gain control circuits 82 via attenuation circuits 81. The output side of each variable gain control circuit 82 is connected to an operational amplifier circuit 83 having an RNF (feedback resistive element) connected between the input and output terminals. These circuits 81 to 83 are formed from bipolar transistors.
The attenuation circuit 81 is made up of an input resistor R1 connected between a signal input node and an attenuation output node, and a first gm amplifier A1 of constant gm type having an equivalent resistor between the attenuation output node and an AC ground node.
The current-controlled variable gain control circuit 82 uses a second gm amplifier A2 of variable gm type in which gm changes in accordance with a control current from the current control circuit 90. By connecting the first gm amplifier A1 to the second gm amplifier A2, as described above, gm variations including the temperature coefficient of the second gm amplifier A2 are cancelled.
To perform tracking error balance adjustment using the tracking error balance adjustment circuit 80 shown in FIG. 2, the tracking error balance adjustment voltage TEB is changed to change an output current (control current) from the current control circuit 90. In accordance with this control current, the operating current of the second gm amplifier A2 is changed (e.g., 50 to 150 xcexcA). At this time, the two circuits are differentially controlled to increase gm of the second gm amplifier A2 in one circuit and decrease gm of the second gm amplifier A2 in the other circuit. Accordingly, the tracking error signal is controlled to 0.
In this case, the control current to one circuit is increased. Then, the operating current of the second gm amplifier A2 increases in proportion to the control current to increase gm and the gain. To the contrary, the control current to the other circuit is decreased. Then, the operating current of the second gm amplifier A2 decreases in proportion to the control current to decrease gm and the gain.
The conventional tracking error balance adjustment circuit 80 suffers the following problems.
The second gm amplifier A2 also functions to drive the output-side circuit RNF. When the control current is decreased to set a small gain, i.e., small gm of the second gm amplifier A2, RNF cannot be satisfactorily driven.
The output offset of the tracking error balance adjustment circuit 80 is desirably set small. For this purpose, gm of the second gm amplifier A2 must be minimized, and a necessary gain and RNF drivability must be ensured.
However, as described above, the control current for controlling gm of the second gm amplifier A2 controls the gain and RNF drivability, which greatly limits the circuit design.
As described above, the tracking error balance adjustment circuit of the conventional optical disk playback apparatus cannot satisfactorily drive the subsequent RNF when gm of the second gm amplifier is set small. If gm of the second gm amplifier A2 is minimized to reduce the output offset, it becomes difficult to ensure a necessary gain and RNF drivability. This greatly limits the circuit design.
The present invention has been made to overcome the conventional drawbacks, and has as its object to provide a tracking error balance adjustment circuit in which the same gain balance adjustment width as in the prior art can be ensured even with a smaller change in control current than in the prior art, gm of a subsequent gm amplifier need not be set so small as to fail to drive an output-stage RNF, and the output offset can be reduced, and an optical disk playback apparatus using the same.
Next, prior art regarding a current control circuit, a variable gain amplifier circuit using the current control circuit, an optical disk playback apparatus using the current control circuit will be explained.
In an optical disk playback apparatus for reading and playing back, with an optical pickup, information data recorded on an optically readable/writable optical disk such as a compact disk, the amplitude of a playback signal varies owing to physical variations (in the reflectance of a reflecting film, the modulation factor, or the like) caused by manufacturing variations in optical disks to be played back.
FIG. 3 schematically shows a conventional optical disk playback apparatus having a playback signal amplitude adjustment function in order to prevent amplitude variations.
In FIG. 3, an optical pickup 61 incorporates a semiconductor laser, photoelectric converter, and the like. The pickup 61 reads digital signals recorded on an optical disk rotated by a motor, and generates an RF (e.g., 1 to 40 MHz) analog signal.
The RF signal output from the pickup 61 is added and amplified by a head amplifier 62. An output signal from the head amplifier 62 is amplified by a variable gain RF amplifier 63 controlled by an AGC (Automatic Gain Control) loop, and at the same time automatically adjusted to have a predetermined amplitude suitable for subsequent signal processing.
An output signal from the variable gain RF amplifier 63 is input to a DSP (digital servo processor) 64. The DSP 64 forms an amplitude adjustment loop for comparing the amplitude of the output signal from the variable gain RF amplifier 63 with a predetermined signal amplitude reference value serving as an adjustment target set in advance, and controlling the gain of the variable gain RF amplifier 63 in accordance with the difference so as to keep the amplitude of the output signal constant.
The output signal from the variable gain RF amplifier 63 is also sent to a binarization circuit (data slice circuit) 65 where the output signal is converted into a binary signal of xe2x80x9cHxe2x80x9d or xe2x80x9cLxe2x80x9d with reference to a predetermined slice level. This binary signal is input to a PLL (Phase-Locked Loop) circuit 66 which generates a clock signal synchronized with the binary signal. A digital signal processing circuit 67 receives the clock signal and binary signal, performs demodulation/error correction, and plays back information data recorded on the optical disk.
Although not shown, this playback apparatus comprises a sliding actuator for sliding the pickup 61 radially along the optical disk, a disk motor for rotating the optical disk at a predetermined rotational speed, and a system controller microcomputer for controlling various servo control circuits.
The amplitude of an AC component (corresponding to information data) included in an output signal from the pickup 61 is influenced by both the modulation factor and the reflectance (depending on a scratch, dirt, and the like on the disk surface) of the reflecting film of an optical disk.
If the modulation factor and the reflectance of an optical disk to be played back are low, the amplitude of an AC component included in the output signal from the pickup 61 is small. In this case, the amplitude adjustment feedback control loop increases the gain of the variable gain RF amplifier 63 in order to control the amplitude of the AC component to a predetermined set value.
To perform accurate binarization in the binarization circuit 65, an output signal from the variable gain RF amplifier 63 must have a proper amplitude. In general, the binarization circuit 65 is designed to execute control of an optimal slice level when an output from the variable gain RF amplifier 63 has a predetermined amplitude.
FIG. 4 shows an arrangement of the variable gain RF amplifier 63 in FIG. 3.
In FIG. 4, the transconductance gm of a gm amplifier 71 is changed by controlling the collector current of an amplifier bipolar transistor.
A feedback resistor 73 is connected between the output terminal of a bipolar type operational amplifier circuit 72 and its inverting input terminal (xe2x88x92) for receiving an output signal from the gm amplifier 71. The non-inverting input terminal (+) of the operational amplifier circuit 72 is connected to a reference potential VRO.
A current control circuit 74 is constituted as follows. Of a differential pair of transistors Q71 and Q72, the base of one transistor Q71 is biased to a proper reference potential VREF, whereas the base of the other transistor Q72 receives a control voltage RFGC from the DSP 64 via a control voltage input terminal 75. The level of an output current is controlled in accordance with the level of the control voltage RFGC, and the output current controls the collector current of the differential amplifier transistor of the gm amplifier 71.
In the current control circuit 74, first and second constant current sources 171 and 172 are respectively connected between a VCC node for receiving a power supply potential VCC, and the emitters of the differential pair of PNP transistors Q71 and Q72. A gain adjustment resistor 76 is connected between these emitters.
The collector of the PNP transistor Q72 is connected to a ground potential GND, whereas the collector of the PNP transistor Q71 is connected to GND via the collector-emitter path of an NPN transistor Q73 having a collector and base connected to each other. The base and emitter of the NPN transistor Q73 are respectively connected (current-mirror-connected) to those of an NPN transistor Q74. The collector of the NPN transistor Q74 is connected to the gain control input node of the gm amplifier 71.
A third constant current source 173 for supplying a predetermined collector current to the NPN transistor Q73 is connected between the VCC node and the collector of the NPN transistor Q73.
The bases of the differential pair of PNP transistors Q71 and Q72 receive the reference potential VREF and control voltage RFGC, respectively. The collector current of the PNP transistor Q72 and that of the NPN transistor Q73 are controlled in accordance with the control voltage RFGC.
More specifically, if the control voltage RFGC exceeds the reference potential VREF, the current of one PNP transistor Q72 out of the differential pair of transistors Q71 and Q72 decreases, and the current of the other PNP transistor Q71 increases.
To the contrary, if the control voltage RFGC becomes lower than the reference potential VREF, the current of one PNP transistor Q72 out of the differential pair of transistors Q71 and Q72 increases, and the current of the other PNP transistor Q71 decreases. In this case, even if the control voltage RFGC drops excessively to turn off the PNP transistor Q71 out of the differential pair of transistors Q71 and Q72, the NPN transistor Q73 receives a predetermined collector current from the third constant current source 173.
The collector current of the NPN transistor Q74 current-mirror-connected to the NPN transistor Q73 controls the magnitude of the collector current of the gm amplifier 71.
FIGS. 5A and 5B respectively show an output current characteristic of the current control circuit 74 in FIG. 4 with respect to the control voltage RFGC and a gain (logarithm expression) characteristic of the gm amplifier 71 in FIG. 4 with respect to the control voltage RFGC.
In the characteristic shown in FIG. 5A, assuming that the output current is 150 xcexcA (IREF) for the control voltage RFGC serving as the reference potential VREF (e.g., 1.65V), the output current is 50 xcexcA (smaller than IREF by 100 xcexcA) for the control voltage RFGC of 0V, and is 250 xcexcA (larger than IREF by 100 xcexcA) for the control voltage RFGC of 3.3V. That is, the current output linearly changes with respect to changes in control voltage RFGC of the current control circuit 74. The change slope is equal between a range where the control voltage RFGC is lower than the reference potential VREF and a range where the control voltage RFGC is higher than the reference potential VREF.
In the characteristic shown in FIG. 5B, assuming that the gain is 0 dB for the control voltage RFGC of 1.65V, the gain is about xe2x88x9210 dB in 20 log(50/150) for the control voltage RFGC of 0V, and is about +4.4 dB in 20 log(250/150) for the control voltage RFGC of 3.3V. That is, the gain of the gm amplifier 71 linearly changes both in the range where the control voltage RFGC of the current control circuit 74 is lower than the reference potential VREF and the range where the control voltage RFGC is higher than the reference potential VREF. However, as for the gain change slope (control sensitivity) of the gm amplifier 71, the control sensitivity is different between the two ranges.
In other words, when the control voltage RFGC increases and decreases equally by 1.65V from the reference potential VREF, the gain change width of the variable gain RF amplifier shown in FIG. 4 becomes different between the ranges higher and lower than the reference potential VREF. This makes it difficult to design the subsequent circuit, resulting in inconvenience.
An output current from the gm amplifier 71 in FIG. 4 functions to drive the feedback resistor 73 between the inverting input terminal (xe2x88x92) and output terminal of the operational amplifier circuit 72. If the gain of the gm amplifier 71 excessively decreases, the level for driving the feedback resistor 73 runs short.
As described above, the variable gain RF amplifier used in the conventional optical disk playback apparatus exhibits different gain control sensitivities of the gm amplifier between the range where the control voltage RFGC input to the current control circuit for controlling the gain of the gm amplifier is lower than the reference potential VREF, and the range where the control voltage RFGC is higher than the reference potential VREF. Even if the control voltage RFGC increases and decreases equally from the reference potential VREF, the gain change width is unbalanced. This makes it difficult to design the subsequent circuit, thereby resulting in inconvenience.
If the gain of the gm amplifier excessively decreases, the level for driving the feedback resistor connected between the input and output terminals of the subsequent operational amplifier circuit runs short.
The present invention has been made to overcome the conventional drawbacks, and has as its object to provide a current control circuit capable of arbitrarily setting the output current so that the sensitivity can be different between the range where the control voltage RFGC is lower than the reference potential VREF and the range where the control voltage RFGC is higher than the reference potential VREF.
It is another object of the present invention to provide a convenient variable gain amplifier circuit in which the a gain control sensitivity of a gm amplifier can be set substantially equal in both of the range where the control voltage RFGC input to the current control circuit is lower than the reference potential VREF and the range where the control voltage RFGC is higher than the reference potential VREF, thereby simplifying the design of the subsequent circuit, and an optical disk playback apparatus using the same.
According to the first aspect of the invention, there is provided A tracking error balance adjustment circuit for correcting at least one of first and second signals in level to set a level difference between the first and second signals to substantially zero when an irradiation position of a laser beam substantially coincides with a track center wherein a tracking error signal is generated in accordance with the level difference between the first and second signals that complementarily change in level in accordance with a deviation of the irradiation position of the laser beam from the track center in order to detect a shift of the irradiation position of the laser beam emitted by an optical pickup from a track which holds information on an optical disk, the tracking error balance adjustment circuit comprising: a current control circuit receiving a tracking error balance adjustment voltage and converting the tracking error balance adjustment voltage into a control current; and a gain control circuit receiving the first signal, the second signal, and the control current, adjusting at least one of the first signal and the second signal in level, and outputting the first signal and the second signal, the gain control circuit comprising at least two signal paths each of which has a plurality of gm amplifiers of variable gm type, the plurality of gm amplifiers in the same signal path have transconductances controlled differentially in accordance with the control current from the current control circuit.
According to the second aspect of the invention, there is provided a semiconductor integrated circuit apparatus employed in an optical disk playback apparatus having a tracking control servo mechanism, the semiconductor integrated circuit apparatus comprising: a tracking error signal generation circuit which generates a tracking error signal, and which includes a tracking error balance adjustment circuit according to the first aspect of the invention, wherein the tracking control servo mechanism controls an irradiation light from an optical pickup in position on an optical disk in the radial direction thereof, in accordance with an output from the tracking error signal generation circuit, there by to control the irradiation light from the optical pickup to be kept on a track.
According to the third aspect of the invention, there is provided A current control circuit comprising: a first circuit receiving a control voltage and a reference potential, and outputting a current in accordance with the control voltage and the reference potential; a second circuit receiving the control voltage and the reference potential, and outputting a current in accordance with the control voltage and the reference potential; and a third circuit outputting a current output generated by adding the output of the first circuit and the output of the second circuit, wherein: the current output changes in accordance with different functions in a range where the control voltage input is lower than the reference potential and a range where the control voltage is not lower than the reference potential.
According to the fourth aspect of the invention, there is provided an optical disk playback apparatus comprising: an optical pickup for reading information data recorded on an optical disk and generating an RF signal corresponding to the information data; a head amplifier for amplifying the output signal from the pickup; and a variable gain amplifier receiving an output signal from the head amplifier, and outputting an output signal with a predetermined amplitude by adjusting the output signal from the head amplifier in amplitude, wherein the variable gain amplifier includes a current control circuit according to the third aspect of the invention.
According to the fifth aspect of the invention, there is provided an equalizing filter for an optical disk playback apparatus, inserted in a signal path of an RF signal generated by an optical pickup for reading information data recorded on an optical disk, the equalizing filter comprising: a first low-pass filter receiving the RF signal on the signal path; a high-pass filter receiving the RF signal on the signal path; an adder circuit receiving output signals from the first low-pass filter and the high-pass filter; and a second low-pass filter receiving an output signal from the adder circuit, wherein: the first low-pass filter, the high-pass filter, and the second low-pass filter each include a variable gain amplifier circuit, the variable gain amplifier circuit having: a current control circuit according to the third aspect of the invention; and a gm amplifier including an amplifier transistor, the transconductance of the gm amplifier varying by controlling a collector current of the amplifier transistor in accordance with the current output from the current control circuit.
According to the sixth aspect of the invention, a current control circuit comprising: a first circuit receiving a control voltage and a reference potential, and outputting a current in accordance with the control voltage and the reference potential; and at least one current generating circuit receiving the control voltage and the reference potential, and outputting a current in accordance with the control voltage and the reference potential, a current output of the current control circuit being generated by adding the current from the at least one current generating circuit to the current from the first circuit, wherein: change characteristic of the current output of the current control circuit follows different functions in respective sections of an expected range of a control voltage input, the range being divided into the respective sections by different reference potentials as inflection points.
Additional objects and advantages of the invention will be set forth in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. The objects and advantages of the invention may be realized and obtained by means of the instrumentalities and combinations particularly pointed out hereinafter.